1. Field of the Invention
The present invention relates to a self-oscillation type switching power supply, and more specifically, to a switching power supply having a high output voltage.
2. Description of the Related Art
A ringing choke converter has been often utilized as a self-oscillation type power supply. FIG. 12 is a circuit diagram of an existing ringing choke converter. In the drawing, numeral 11 represents a DC power-supply circuit for generating about 120 V DC voltage by rectifying and smoothing a commercial AC power supply and T represents a transformer having a primary winding L.sub.p, a secondary winding L.sub.s, and a feedback winding L.sub.f. Q1 is a switching transistor connected to the DC power-supply through the primary winding L.sub.p of the transformer. To the base of the switching transistor Q1 a starting resistor R1 is connected. Between the feedback winding L.sub.f and the base of the switching transistor Q1 a current-limiting resistor R2, a speed-up capacitor C2, and a diode D2 are connected. Further, between the base and emitter of the switching transistor Q1 a controlling transistor Q2 is connected and across the feedback winding L.sub.f a time constant circuit 4 comprising a resistor R5 and a capacitor C3 is arranged, and they are connected so that the voltage across the capacitor C3 is applied to the base of the controlling transistor Q2. To the secondary winding Ls a rectifying and smoothing circuit 2 comprising a rectifier diode D1 and a smoothing capacitor C1 is connected. To the output side of the rectifying and smoothing circuit 2 a voltage-dividing resistor circuit comprising a resistor R3 and resistor R4, a variable shunt regulator 12, and a photo coupler are connected. The photo-transistor in this photo coupler PC is connected in the charging path for the capacitor C3.
The operation of the power supply shown in FIG. 12 is as in the following. When a DC voltage is applied from the DC power-supply circuit 11, a very small amount of starting current flows into the base of the switching transistor Q1 through the starting resistor R1. As a result, a current flows through the collector of the transistor Q1 and the voltage between the collector and emitter reduces. Then, a voltage is applied across the terminals of the primary winding L.sub.p of the transformer T and an induced voltage proportional to the applied voltage is generated across the feedback winding L.sub.f. Because of this induced voltage, a positive feedback current is supplied to the base of the switching transistor Q1 through the current-limiting resistor R2, and the speed-up capacitor C2 and the diode D2 in parallel, and the transistor Q1 is switched to the ON state (saturation). When the transistor Q1 is turned on, a DC voltage is applied between the terminals of the primary winding L.sub.p of the transformer T and a current flows through the primary winding L.sub.p to cause excitation of the transformer T. At this time, the induced voltage generated across the feedback winding L.sub.f at the same time charges the capacitor C3 through the resistor R5, speed-up capacitor C2, diode D2, and photo-transistor of the photo coupler PC. When the charged voltage of this capacitor C3 reaches the threshold voltage (about 0.6 V) between the base and emitter of the controlling transistor Q2, the base and emitter of the switching transistor Q1 is short circuited. This cuts off the base current of the switching transistor Q1 to turn off the transistor Q1 rapidly. Here, the duration in which the switching transistor Q1 is turned on is equal to the time from start of charging the capacitor C3 to attainment of the voltage across the capacitor C3 to about 0.6 V. When the switching transistor Q1 is turned off, the base of the switching transistor Q1 is reverse biased to a negative potential by the induced voltage across the feedback winding L.sub.f. At the same time the electric charge of the capacitor C3 is forced to discharge (reverse charge) by the feedback winding L.sub.f through the resistor R5, and so the base of the controlling transistor Q2 is reverse biased to a negative potential. Therefore, until the excitation energy of the transformer T is fully discharged through the secondary winding Ls, the switching transistor Q1 continues to be turned off. When the excitation energy of the transformer T has been fully discharged, the induced voltage of the feedback winding L.sub.f disappears rapidly, but because of the leakage induction and distributed capacitance of the transformer T a ringing voltage (kick voltage) is generated so as to forward bias the base of the switching transistor Q1 and the switching transistor Q1 is turned on again. After that, the above-mentioned turn-on and turn-off operation is repeated so that the oscillation grows and continues.
Here, when V.sub.out represents an output voltage of the rectifying and smoothing circuit 2, I.sub.out a current flowing in the load, Lp an inductance of the primary winding L.sub.p, and I.sub.cp the peak value of the collector current of the switching transistor Q1, the output voltage Vout can be given by the following approximate expression. EQU V.sub.out =(L.sub.p .multidot.I.sub.cp.sup.2)/(2 I.sub.out) (1)
Further, when t.sub.on represents a turn-on time of the switching transistor Q1, and Vin a voltage applied between the terminals of the primary winding L.sub.p, then I.sub.cp is given by the following expression. EQU I.sub.cp =(Vin/L.sub.p) t.sub.on (2)
From the relation shown by the expressions (1), (2), by detecting the output voltage and adjusting the current of the photo-transistor in the photo coupler PC and by controlling the turn-on time of the switching transistor Q1, the output voltage V.sub.out can be kept constant.
Then, in the conventional self-oscillation type switching power supply shown in FIG. 12, the output voltage is, for example, as low as 5 V, and the transformer is a step-down transformer. In the construction of the conventional power supply shown in FIG. 12, if the turn ratio of the secondary winding Ls to the primary winding L.sub.p of the transformer T is raised, a power supply for generating a high voltage can be theoretically constructed, but the following problems arise.
FIG. 13 is a circuit diagram of a transformer. C.sub.s represents a distributed capacitance between the terminals of the secondary winding Ls and C.sub.ps represents a distributed capacitance generated between the primary winding L.sub.p and secondary winding Ls. Further, C.sub.pp represents the distributed capacitance C.sub.s and C.sub.ps changed as a capacitance between the terminals of the primary winding L.sub.p. For example, in an electrophotographic copier and page printer a power supply to multiply a DC input voltage of tens V to a DC or AC voltage of hundreds to thousands V is required, but in order to realize such a performance the turn ratio of the secondary winding Ls to the primary winding L.sub.p must be increased to a great extent. Here, when N.sub.p represents the number of turns of the primary winding L.sub.p, Ns the number of turns of the secondary winding L.sub.s, C.sub.s and C.sub.ps the value of distributed capacitance C.sub.s and C.sub.ps respectively, a distributed capacitance C.sub.pp changed as a capacitance between the terminals of the primary winding L.sub.p is given by the following approximate expression. EQU C.sub.pp =(C.sub.s +C.sub.ps).times.(N.sub.s /N.sub.p).sup.2 (3)
Accordingly, the capacitance C.sub.pp in a high-voltage transformer becomes extremely large compared with that in a low-voltage transformer.
When the transformer T in FIG. 12 is replaced with the high-voltage transformer shown in FIG. 13, the primary converted distributed capacitance C.sub.pp of the transformer becomes extremely large compared with that of the low-voltage transformer. Therefore, when the switching transistor is turned on, excess current flows and switching loss is increased, and also the amplitude of the ringing component superposed on the collector current of the switching transistor is increased. As a result, the output voltage control or output current control is adversely affected. This is caused by the fact that, as shown in the expression (3), the distributed capacitance C.sub.pp converted as the capacitance between the terminals of the primary winding becomes extremely large.
Here, an equivalent circuit of the high-voltage transformer is shown in FIG. 14. In the drawing L1 and L2 represents a leakage inductance, L.sub.p an inductance of the primary winding, and C.sub.pp a primary converted distributed capacitance. FIG. 15 is the diagram of a waveform showing the collector current of the switching transistor Q1 and others. FIG. 16 is the diagram of the waveform showing the relation between the collector voltage and current and between the base voltage and current of the switching transistor Q1. In FIG. 15 V.sub.Lp represents the waveform of the applied voltage to the primary winding, I.sub.c ' the current flowing in the primary converted distributed capacitance C.sub.pp, I.sub.Lp the current flowing in the primary winding, and I.sub.c the collector current of the switching transistor Q1. When the switching transistor Q1 is turned on, the current I.sub.c ' flows through the capacitance C.sub.pp in such a way that initially excess current to charge the capacitance C.sub.pp flows and after completion of the charge the capacitance C.sub.pp becomes resonant with the leakage inductance (L1, L2) and the oscillation is repeatedly damped. The combined current of I.sub.c ' and I.sub.Lp flows through the collector of the switching transistor Q1, and accordingly as shown by I.sub.c in FIG. 15 excess current flows initially and then the current with ringing is increased at a slope of V.sub.Lp /L.sub.p. At this time, the ringing component superposed on the collector current I.sub.c has adverse effects of inducing intermittent operation and others on the controlling method to stabilize the output by adjustment of the turn-on time of the switching transistor Q1.
Further, in FIG. 16 V.sub.ce represents a voltage between the collector and emitter of the switching transistor Q1, I.sub.c a collector current of the transistor Q1, V.sub.be a voltage between the base and emitter of the transistor Q1, and I.sub.b a base current of the transistor Q1. Because of excess current flowing in the collector of the switching transistor Q1 when the transistor Q1 is turned on, as shown by the hatched area in FIG. 16 (a), large switching loss is caused by the product of the collector current and the voltage between the collector and emitter at the time when the switching transistor Q1 is turned on.
In order to solve the above-mentioned problems, in conventional high-voltage power supplies, as shown in FIG. 17 it was common to stabilize an output by adjusting a DC input voltage to be input to the primary winding of a high-voltage transformer, not by adjusting the turn-on time of a switching transistor Q1. In FIG. 17 a transistor Q5 is a controlling power transistor to step down the voltage of a DC input power supply 1 and adjust the input voltage to a high-voltage transformer T. A control circuit stabilizes an output voltage by controlling the base current of the transistor Q5 in accordance with a detected signal from an output voltage detection circuit. The switching transistor Q1 is always turned on and off at a constant cycle by an oscillation circuit.
However, in the circuit shown in FIG. 17, a construction of a separate excitation type switching power supply is required, and accordingly an oscillation circuit is externally required and in addition a power transistor to step down the input voltage to the high-voltage transformer is separately required. As a result, the circuit construction becomes complicated and large-sized.